Radio-frequency (RF) receiver circuit architectures are typically designed to convert radiated signals at very high frequencies (typically referred to as radio frequencies) to baseband signals centered around zero frequency from which digital information may be extracted. These architectures typically interface to an antenna, which converts the radiated signal into a current or voltage, which is then processed by the RF receiver circuitry such that it is of the form (proper signal amplitude, center frequency, and bandwidth) that may interface to an analog-to-digital converter (ADC), which converts the processed analog signal into a form that may be further manipulated by digital circuit blocks such as microprocessors.
Typically, the RF receiver is only interested in extracting signal content from a much smaller frequency range than is acquired by the antenna. In order to avoid overloading the analog-to-digital converter at the terminus of the RF receiver, the undesired frequency bands must be filtered out. The cost of a filter in general is proportional to the product of the filter selectivity and the filter center frequency. Thus, completely removing the signals from the unwanted frequency range in the first few stages of the RF receiver, where the signal center frequency is quite high, is infeasible. Typically, one or more expensive off-chip filters are used to partially attenuate the undesired frequency bands within the first few blocks of the RF receiver. After sufficient gain is achieved in the desired signal band to surmount the noise introduced by later stages, a mixer is employed in order to perform a linear frequency translation of the desired signal to a lower frequency such that low cost, high selectivity filtering can be performed monolithically with the rest of the circuit.
These signals in the unwanted frequency range typically interact with nonlinearities within the RF circuitry and may produce undesirable signal products within the desired signal band. The undesirable signal products in question typically arise from the third-order terms in the circuit nonlinearities and are hence called in the literature third-order intermodulation products, or IM3 products for short. These interactions would typically occur in the first few blocks of the RF receive chain, prior to the bulk of selective filtering in the system. If either the costly filters mentioned above were removed to reduce component count or the block nonlinearities were reduced in order to save power or area, the undesirable signal products can have the effect of disabling the operation of the complete RF receiver system.
There exists a significant economic incentive to reduce the external component count, power, and area of RF receivers. If an efficient way to improve the linearity of an RF receiver could be devised, its power and area could be reduced while still retaining an acceptable possibility of system failure. If the linearity of an RF receiver were improved further to the point where certain thresholds were met, the off-chip filters could also be removed with an acceptable possibility of system failure.
Given that nonlinear transfer functions are inherently associated with the amplification components of contemporary RF receivers, one way to improve the nonlinearity of the system while only adding a minimal power burden is to provide accurate cancellation of the generated IM3 products.
The amplitude of the IM3 products of a receiver typically increases as the cube of the amplitude of the input signal to the receiver. The input amplitude at which the desired output amplitude and the rms amplitude of the IM3 products are equal (this is typically an extrapolated value) is referred to as the input-intercept-3 point (IIP3) and is a common measure of both receiver and receiver block linearity.
The most well known technique to improve the IIP3 of a given block in the RF receiver (such as the LNA or mixer) is to use emitter or source degeneration in the transconductance amplifiers in order to shift some of the transconductance burden from a nonlinear semiconductor junction to an inherently linear passive component, such as a resistor or an inductor. The use of a resistor for degeneration purposes, however, increases the noise contributed by the block. Typically the desired signal amplitude is small and any additional noise contributions can easily bury the desired signal.
This issue can be avoided by degenerating the transconductance amplifier with an inductor; however, this solution requires a large amount of area and is frequency-selective, that is, it would be unsuitable for emerging wideband communication systems such as orthogonal frequency-division multiplexing (OFDM). Furthermore, the level of improvement obtained is limited and has the effect of reducing the gain and other desirable properties of the amplifier.
In van der Heijden, M. P. et al., “A Novel Frequency-Independent Third-Order Intermodulation Distortion Cancellation Technique for BJT Amplifiers,” Solid-State Circuits Conference, IEEE Journal of, Volume 37, Issue 9, September 2002 Page(s): 1176-1183, a Volterra-series analysis is performed on a BJT common-emitter amplifier. This paper brings up the point that in order to completely cancel IM3 products in a typical single-ended BJT common-emitter amplifier, unacceptable conditions must be imposed on the input and output loading on the amplifier from a noise and power match perspective.
A differential BJT common-emitter amplifier with more degrees of freedom is introduced and analyzed in the paper using the same Volterra series analysis. It is found that under certain assumptions, IM3 product cancellation can be achieved while maintaining optimal input and output loading on the amplifier.
In general, bias-point specific compensation techniques such as these are sensitive to process, supply voltage, and temperature when implemented monolithically on a silicon die. For this technique to work optimally, an external monitoring and bias adjustment circuit would be required. Also, measurements of the actual circuit reported in this paper show a 30 dB reduction in peak IIP3 from the idealized circuit in simulation, suggesting that the cancellation technique is not robust in practice.
Furthermore, the differential nature of this amplifier is problematic in that it requires two on-chip center-tapped transformers for a monolithic implementation. In practice, this technique is area-intensive and would seldom be incorporated in a commercial integrated circuit.
In Aparin, V. et al., “Modified Derivative Superposition Method For Linearizing FET Low Noise Amplifiers,” Radio Frequency Integrated Circuits (RFIC) Symposium. 2004. Digest of Papers. 2004 IEEE, 6-8 Jun. 2004 Page(s) 105-108 (Aparin et al. I) and Tae Wook Kim et al., “Highly Linear Receiver Front-End Adopting MOSFET Transconductance Linearization By Multiple Gated Transistors,” Solid-State Circuits Conference, IEEE Journal of, Volume 39, Issue 1, January 2004 Page(s):223-229, a derivative superposition technique is used to linearize the low noise amplifier (LNA) and the LNA and mixer, respectively, in an RF receive chain. Both references implement this technique in an economically competitive CMOS process as opposed to a relatively expensive BJT process.
This technique relies on the fact that the coefficient of the 3rd-order term in the Taylor series expansion for the transconductance of a MOS common-source amplifier is bias-dependent. Thus, two MOS common source amplifiers biased at different operating points can be placed in parallel, one main amplifier biased at the optimal bias point for gain and noise purposes, and one auxiliary amplifier biased at the proper bias point to generate IM3 terms at the same amplitude but opposite polarity as the main amplifier.
This technique is power-efficient since the desirable bias point for the auxiliary amplifier is such that it is in weak inversion; that is, it dissipates negligible quiescent current.
Aparin, V. et al. I and Tae Wook Kim et al. describe additional modifications of this technique to improve performance in the presence of harmonic feedback, an effect originally discounted during the inception of this technique.
In addition, Tae Wook Kim et al. show that with proper biasing, the best-case IIP3 drops by only about 4 dBm in the case of the LNA, showing that the performance does not change drastically with reasonable changes in process, voltage, and temperature.
However, because this technique depends on the gate biasing of the circuit, the beneficial effects of this technique will break down for large input voltage swings, which is especially a problem at the mixer. This is supported in the paper by Tae Wook Kim et al., where the IIP3 curve is not a straight line.
In some designs, such as by Tae Wook Kim et al., the auxiliary transistor has been designed to be a similar size as the main transistor. In the event of a large signal swing, both transistors will conduct heavily, leading to increased large-signal current for a large input signal. This reduces the compression point (ICP1) of the circuit and hence increases its susceptibility to desensing by extremely large signals. Furthermore, the mixer reported in Tae Wook Kim et al. (no mixer improvement was undertaken using this technique in the paper by Aparin, V. et al.) only achieves an IIP3 of at most +9 dBm (+7 dBm at least). According to Aparin, V. et al., “Integrated LMS Adaptive Filter of TX Leakage for CDMA Receiver Front Ends,” Solid-State Circuits Conference, IEEE Journal of, Volume 41, Issue 5, May 2006 Page(s):1171-1182, (Aparin, V. et al. II) in order to justify the removal of an off-chip filter in CDMA a mixer IIP3 of +24 dBm is required. Hence, further improvement is desirable.
In CDMA and WCDMA the dominant IM3 producer is the leakage from the transmit (Tx) signal through the front-end duplexer. This leakage, which can be as high as −22 dBm at the receiver LNA, interacts with other unwanted out-of-band signals to occasionally produce in-band IM3 products.
In Aparin, V. et al. II, a separate leakage path is introduced directly from the transmit path to the receive path. This leakage signal is subtracted from the duplexer leakage signal, thus removing it. With this signal gone, the linearity requirements of the receiver are significantly reduced. All of the above-mentioned approaches appear to provide circuits that are purely analog in nature.
Since the amount of duplexer leakage varies based on a number of factors, including process, temperature, and duplexer variations, the amount of leakage is controlled by an LMS algorithm implemented in analog circuitry.
However, the LMS algorithm does not in this case compensate for delay through the duplexer and its performance is thus limited over chip-to-chip variation to less than what is required to remove the off-chip filter prior to the mixer.
The LMS filter also substantially increases the front-end noise of the RF receiver (Noise Figure increases from 1.6 dB to almost 3 dB.)
Therefore, there is a need in the art to provide efficient and accurate IM3 cancellation in RF receivers.